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19-0796; Rev 0; 4/07
Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response
General Description
The MAX8664 dual-output PWM controller is a low-cost, high-performance solution for systems requiring dual power supplies. It provides two individual outputs that operate 180 out-of-phase to minimize input current ripple, and therefore, capacitance requirements. Built-in drivers are capable of driving external MOSFETs to deliver up to 25A output current from each channel. The MAX8664 operates from a 4.5V to 28V input voltage source and generates output voltages from 0.6V up to 90% of the input voltage on each channel. Total output regulation error is less than 0.8% over load, line, and temperature. The MAX8664 operates with a constant switching frequency adjustable from 100kHz to 1MHz. Built-in boost diodes reduce external component count. Digital softstart eliminates input inrush current during startup. The second output has an optional external REFIN2, facilitating tracking supply applications. Each output is capable of sourcing and sinking current, making the device a great solution for DDR applications. The MAX8664 employs Maxim's proprietary peak voltage-mode control architecture that provides superior transient response during either load or line transients. This architecture is easily stabilized using two resistors and one capacitor for any type of output capacitors. Fast transient response requires less output capacitance, consequently reducing total system cost. The MAX8664B latches off both controllers during a fault condition, while the MAX8664A allows one controller to continue to function when there is a fault in the other controller.
Features
o 0.8% Output Accuracy Over Load and Line o Operates from a Single 4.5V to 28V Supply o Simple Compensation for Any Type of Output Capacitor o Internal 6.5V Regulator for Gate Drive o Integrated Boost Diodes o Adjustable Output from 0.6V to 0.9 x VIN o Digital Soft-Start Reduces Inrush Current o 100kHz to 1MHz Adjustable Switching o 180 Out-of-Phase Operation Reduces Input Ripple Current o Overcurrent and Overvoltage Protection o External Reference Input for Second Controller o Prebiased Startup Operation
MAX8664
Ordering Information
PART MAX8664AEEP+ MAX8664BEEP+ PINPACKAGE 20 QSOP 20 QSOP PKG CODE E20-1 E20-1 FAULT ACTION Independent Joint
Note: This device operates over the -40C to +85C operating temperature range.
+Denotes lead-free package.
Typical Operating Circuit
IN2 OUT2 IN ILIM2 DH2 BST2 LX2 DL2 VL ILIM1 DH1 BST1 LX1 IN1 OUT1
Applications
Desktop and Notebook PCs Graphic Cards ASIC/CPU/DSP Power Supplies Set-Top Box Power Supply Printer Power Supply Network Power Supply POL Power Supply
DL1
MAX8664
GND FB2 REFIN2 VCC PGND FB1 PWRGD OSC/EN12
Pin Configuration appears at end of data sheet.
________________________________________________________________ Maxim Integrated Products
1
For pricing delivery, and ordering information please contact Maxim Direct at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
ABSOLUTE MAXIMUM RATINGS
IN to GND ..............................................................-0.3V to +30V VL to GND...................................................................-0.3 to +8V IN, BST_ to VL ........................................................-0.3V to +30V VCC, FB_, PWRGD to GND.......................................-0.3V to +6V VL to VCC ....................................................................-2V to +8V PGND to GND .......................................................-0.3V to +0.6V DL_ to PGND...............................................-0.3V to (VVL + 0.3V) DH_ to PGND............................................-0.3V to (VBST_+ 0.3V) BST_ to GND.............................................................-0.3V to 38V BST_ to LX ................................................................-0.3V to +8V LX_ to PGND .................-1V (-2.5V for < 50ns transient) to +30V DH_ to LX_................................................-0.3V to (VBST_+ 0.3V) Note 1: Package mounted on a multilayer PCB. ILIM_ to GND ...............................................-0.3V to (VIN + 0.3V) ILIM_ to LX_............................................................-0.6V to +30V OSC/EN12, REFIN2 to GND .....................-0.3V to (VVCC + 0.3V) VL Continuous Current ..............................................125mARMS VCC Continuous Current..............................................10mARMS Continuous Power Dissipation (TA = +70C) (Note 1) 20-Pin QSOP (derate 11.0mW/C above +70C).........884mW Operating Temperature Range ...........................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = 12V, ROSC/EN12 to GND = 56.1k, REFIN2 = VCC, TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C.) (Note 2)
PARAMETER SUPPLY VOLTAGES IN Supply Voltage VL Output Voltage VCC Output Voltage VCC Undervoltage Lockout (UVLO) Standby Supply Current Operating Supply Current REGULATOR SPECIFICATIONS Reference Accuracy TA = 0C to +85C TA = -40C to +85C VREFIN2 = VVCC VREFIN2 = 1.000V REFIN2 to Internal Reference Switchover Threshold REFIN2 Maximum Program Voltage REFIN2 Disable Threshold FB Input Bias Current REFIN2 Bias Current FB Propagation Delay VFB = 0.5V VREFIN2 = 0.65V FB rising to DH falling Not to be switched during operation TA = 0C to +85C TA = -40C to +85C 0.5955 0.5930 0.5952 0.5925 0.995 2 0.600 0.600 0.600 0.600 1.000 VVCC 0.7 1.3 50 3 3 90 0.6045 0.6070 0.6048 0.6075 1.005 VVCC 0.3 V V mV nA nA ns V V 7.2 IN = VL = VCC 7.2V < VIN < 28V, 0 < IVL < 60mA 7.2V < VIN < 28V, 0 < ICC < 5mA Rising Hysteresis OSC/EN12 not connected No switching, VFB_ = 0.65V VIN = 12V, IIN VCC = VIN = VVL = 5V, IIN + IVL + IVCC VIN = 12V, IIN VCC = VIN = VVL = 5V, IIN + IVL+ IVCC 4.5 6.10 4.5 3.4 6.6 5.0 3.5 350 0.095 0.08 1.4 1.1 0.2 0.2 2.5 1.8 28.0 5.5 6.75 5.5 3.6 V V V V mV mA mA CONDITIONS MIN TYP MAX UNITS
FB_ Regulation Accuracy
2
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 12V, ROSC/EN12 to GND = 56.1k, REFIN2 = VCC, TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C.) (Note 2)
PARAMETER PROTECTION FEATURES Overvoltage Protection (OVP) Threshold VFB1 rising VFB2 rising, VREFIN2 1.3V VREFIN2 = VVCC, VFB_ rising, MAX8664B Power-Good (PWRGD) Threshold High-Side Current-Sense Program Current (Note 3) ILIM Leakage High-Side Current-Sense Overcurrent Trip Adjustment Range Internal Soft-Start Time REFIN2 Internal Pulldown Resistance Thermal-Shutdown Threshold DRIVER SPECIFICATIONS Sourcing current, IDH = -50mA Sinking current, IDH = 50mA Sourcing current, IDL = -50mA Sinking current, IDL = 50mA Dead Time for Low-Side to High-Side Transition DH_ Minimum On-Time BST Current Internal Boost Switch Resistance PWM CLOCK OSCILLATOR PWM Clock-Frequency Accuracy PWM Clock-Frequency Adjustment Range OSC/EN12 Disable Current ROSC/EN12 = 226k to 22.6k -15 100 1.5 +15 1000 2.5 % kHz A VBST - VLX = 7V, VLX = 28V, VFB_ = 0.55V OSC/EN12 not connected DL_ falling to DH_ rising VVL = 6.5V VIN = VVL = VVCC = 5V VVL = 6.5V VIN = VVL = VVCC = 5V VVL = 6.5V VIN = VVL = VVCC = 5V VVL = 6.5V VIN = VVL = VVCC = 5V VVL = 6.5V VVL = 5V 70 13 1.35 1.55 0.9 1.0 1.3 1.5 0.6 0.7 25 28 108 1.25 0.001 6 149 2.3 43 ns ns mA A 1.1 2 1.4 2.1 ROSC/EN12 = 56.1k, 400kHz Engaged momentarily at startup Junction temperature VFB1 rising, MAX8664A Hysteresis TA = +85 C TA = +25oC TA = +25C TA = +85C 0.05 2.5 10 +160 20 44
o
MAX8664
CONDITIONS
MIN
TYP 0.75 REFIN2 + 0.15
MAX
UNITS
V
0.500
0.525 5 60 50 0.1 0.1
0.550
V %
60 1.0
A A V ms C
0.40
DH_ Driver Resistance
DL_ Driver Resistance
Note 2: Specifications at -40C are guaranteed by design and not production tested. Note 3: This current linearly compensates for the MOSFET temperature coefficient.
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
Typical Operating Characteristics
(Circuit of Figure 2, 600kHz, VIN = 12V, VOUT1 = 2.5V, VOUT2 = 1.8V, TA = +25C, unless otherwise noted.)
EFFICIENCY vs. LOAD CURRENT (600kHz, FIGURE 2)
MAX8664 toc01
EFFICIENCY vs. LOAD CURRENT (1MHz, FIGURE 4)
MAX8664 toc02
LOAD REGULATION (600kHz, FIGURE 2)
2.54 2.53 OUT1 VOLTAGE (%) 2.52 2.51 2.50 2.49 2.48 2.47 2.46 2.45 0 2 4 6 8 10 OUT1 LOAD CURRENT (A) IOUT2 = 0A IOUT2 = 8A IOUT2 = 4A
MAX8664 toc03
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.1 1 LOAD CURRENT (A) NO LOAD ON THE OTHER REGULATOR VOUT = 1.8V VOUT = 2.5V
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 VIN = 3.3V VVL = 5V NO LOAD ON OUT2 0.1 1 LOAD CURRENT (A) VOUT1 = 1.8V VOUT1 = 2.5V
2.55
10
10
LINE REGULATION (600kHz, FIGURE 2)
2.54 2.53 OUT1 VOLTAGE (%) 2.52 2.51 2.50 2.49 2.48 2.47 2.46 2.45 6 8 10 12 14 16 18 20 INPUT VOLTAGE (V) 0 NO LOAD 50 8A LOAD
MAX8664 toc04
ROSC/EN12 vs. SWITCHING FRQUENCY
MAX8664 toc05
OUT1 LOAD TRANSIENT (FIGURE 2)
MAX8664 toc06
2.55
250
200 ROSC/EN12 (k)
150
VOUT2
100mV/div
100 IOUT2 2.5A
5A
2.5A
2A/div
100
400 700 SWITCHING FREQUENCY (kHz)
1000
20s/div
LOAD TRANSIENT -3A TO +3A TO -3A (FIGURE 3)
MAX8664 toc07
POWER-UP WAVEFORMS
MAX8664 toc08
10V/div VOUT1 50mV/div VIN 2V/div VOUT2 +3A IOUT2 -3A -3A 5A/div VPRWGD 100s/div 1ms/div 50mV/div VOUT1 VOUT2 5V/div 2V/div
4
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response
Typical Operating Characteristics (continued)
(Circuit of Figure 2, 600kHz, VIN = 12V, VOUT1 = 2.5V, VOUT2 = 1.8V, TA = +25C, unless otherwise noted.)
MAX8664
POWER-DOWN WAVEFORMS
MAX8664 toc09
ENABLE WAVEFORMS (FIGURE 2)
MAX8664 toc10
VIN 10V/div VOUT1 VOUT2 2V/div 2V/div VPRWGD 5V/div 1ms/div VPRWGD 1ms/div 5V/div VOUT1 VOUT2 2V/div 2V/div ENABLE 5V/div
ENABLE WAVEFORMS (FIGURE 4)
MAX8664 toc11
SWITCHING WAVEFORMS
MAX8664 toc12
FEEDBACK VOLTAGE vs. TEMPERATURE
604
MAX8664 toc13
605 10V/div FEEDBACK VOLTAGE (mV)
ENABLE
5V/div
VLX1
603 602 601 600 599 598 597
IL1 VOUT1 VOUT2 1V/div 1V/div VLX2 IL2 2s/div
5A/div
10V/div
VPRWGD 400s/div
5V/div
5A/div
596 595 -40 -20 0 20 60 40 TEMPERATURE (C)
NO LOAD 80 100
SHORT-CIRCUIT WAVEFORMS
MAX8664 toc14
OVERVOLTAGE PROTECTION
MAX8664 toc15
VOUT1 VOUT1 2V/div IIN 2A/div IL1 VDH1 VPRWGD 5A/div 5V/div 10s/div 20s/div VDL1 10V/div 10V/div IL1 10A/div 5V/div
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
Pin Description
PIN 1 NAME DH1 FUNCTION High-Side MOSFET Driver Output for Controller 1. Connect DH1 to the gate of the high-side MOSFET. DH1 is low in shutdown and UVLO. External Inductor Connection for Controller 1. Connect LX1 to the switching node of the MOSFETs and inductor. Make sure LX1 is close to the source of the high-side MOSFET(s) to form a Kelvin connection for high-side current sensing. LX1 is high impedance during monotonic startup and shutdown. Boost Capacitor Connection for the High-Side MOSFET Driver for Controller 1. Connect a 0.22F ceramic capacitor from BST1 to LX1. Low-Side MOSFET Driver Output for Controller 1. Connect DL1 to the gate of the low-side MOSFET(s) for controller 1. DL1 is low in shutdown and UVLO. Low-Side Gate Drive Supply and Output of the 6.5V Linear Regulator. Connect a 4.7F ceramic capacitor from VL to PGND. When using a 4.5V to 5.5V supply, connect VL to IN. VL is the input to the VCC supply. Do not load VL when IC is disabled. Power Ground. Connect to the power ground plane. Connect power and analog grounds at a single point near the output capacitor's ground. Low-Side MOSFET Driver Output for Controller 2. Connect DL2 to the gate of the low-side MOSFET(s) for controller 2. DL2 is low in shutdown and UVLO. Boost Capacitor Connection for the High-Side MOSFET Driver for Controller 2. Connect a 0.22F ceramic capacitor from BST2 to LX2. External Inductor Connection for Controller 2. Connect LX2 to the switching node of the MOSFETs and inductor. Make sure LX2 is close to the source of the high-side MOSFET(s) to form a Kelvin connection for high-side current sensing. LX2 is high impedance during monotonic startup and shutdown. High-Side MOSFET Driver Output for Controller 2. Connect DH2 to the gate of the high-side MOSFET(s) for controller 2. DH2 is low in shutdown and UVLO. Current-Limit Set for Controller 2. Connect a resistor from the drain of the high-side MOSFET(s) to ILIM2. See the Setting the Overcurrent Threshold section. Feedback Input for Controller 2. Connect FB2 to the center of a resistor-divider connected between the output of controller 2 and GND to set the desired output voltage. VFB2 regulates to VREFIN2 or the internal 0.6V reference. To use the internal reference, connect REFIN2 to VCC. External Reference Input for Controller 2. To use the internal 0.6V reference, connect REFIN2 to VCC. To use an external reference, connect REFIN2 through a resistor (> 1k) to a reference voltage between 0V and 1.3V. An RC lowpass filter is recommended when using an external reference and soft-start is not provided by the external reference. For tracking applications, connect REFIN2 to the center of a resistor voltage-divider between the output of controller 1 and GND (see Figure 3). Connect REFIN2 to GND to disable controller 2.
2
LX1
3 4
BST1 DL1
5
VL
6 7 8
PGND DL2 BST2
9
LX2
10 11
DH2 ILIM2
12
FB2
13
REFIN2
14
Switching Frequency Set Input. Connect a 22.6k to 226k resistor from OSC/EN12 to GND to set the switching frequency between 1000kHz and 100kHz. Connect a switch in series with this resistor for OSC/EN12 enable/shutdown control. When the switch is open, the IC enters low-power shutdown mode. In shutdown, OSC/EN12 is internally driven to approximately 800mV. IN GND Internal 6.5V Linear Regulator Input. Connect IN to a 7.2V to 28V supply, and connect a 0.47F or larger ceramic capacitor from IN to PGND. When using a 4.5V to 5.5V supply, connect IN to VL. Analog Ground. Connect to the analog ground plane. Connect the analog and power ground planes at a single point near the output capacitor's ground.
15 16
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response
Pin Description (continued)
PIN 17 NAME VCC FUNCTION Internal Analog Supply. VCC regulates to 1.5V below VVL. Connect a 1F ceramic capacitor from VCC to GND. When using a 4.5V to 5.5V supply, connect a 10 resistor from VCC to IN. VCC is used to power the IC's internal circuitry. Open-Drain Power-Good Output. PWRGD is high impedance when controllers 1 and 2 (using the internal reference) are in regulation. PWRGD is low if the outputs are out of regulation, if there is a fault condition, or if the IC is shut down. PWRGD does not reflect the status of output 2 in the MAX8664A or when REFIN2 is connected to an external reference in the MAX8664B. Feedback Input for Controller 1. Connect FB1 to the center of a resistor-divider connected between the output of controller 1 and GND to set the desired output voltage. VFB1 regulates to 0.6V. Current-Limit Set for Controller 1. Connect a resistor from the drain of the high-side MOSFET(s) to ILIM1. See the Setting the Overcurrent Threshold section.
MAX8664
18
PWRGD
19 20
FB1 ILIM1
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
VCC CURRENT-LIMIT COMPARATOR BIAS GENERATOR UVLO CIRCUITRY 50A LX1 VOLTAGE REFERENCE REF SOFT-START 1 EN SHUTDOWN CONTROL LOGIC THERMAL EN SHUTDOWN DH1 LX1 CONTROL LOGIC
ILIM1 BST1 BST CAP CHARGING SWITCH
REF
SHUTDOWN 1
SHUTDOWN 2 DL1 CLOCK 1 SHUTDOWN 1 PGND VL ILIM2 BST2 BST CAP CHARGING SWITCH
FB1
PWM COMPARATOR 1 0.6V
CURRENT-LIMIT COMPARATOR 50A LX2
S2 FB2 REF2 LX2 S1 ENABLE2 REFIN2 50mV IF VREFIN2 > 2.0V OPEN S1 AND CLOSE S2. OTHERWISE, CLOSE S1 AND OPEN S2. CLOCK 2 SHUTDOWN 2 SOFT-START CLOCK 1 OSCILLATOR CLOCK 2 ENABLE OSC/EN12 DL2 CONTROL LOGIC PWM COMPARATOR 2
DH2
REF
THERMAL SHUTDOWN VL
THERMAL SHUTDOWN
FB1 REF1 - 0.1V
4A
PWRGD
IN
6.5V LDO 1.5V
FB2 REF2 - 0.1V VCC GND
Figure 1. Functional Diagram
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response
Detailed Description
The MAX8664 dual-output PWM controller is a low-cost solution for dual power-supply systems. It provides two individual outputs that operate 180 out-of-phase to minimize input capacitance requirements. Built-in drivers are capable of driving external MOSFETs to deliver up to 25A of current from each output. The MAX8664 operates from a 4.5V to a 5.5V or a 7.2V to 28V input and generates output voltages from 0.6V up to 90% of the input voltage on each channel. Total output error is less than 0.8% over load, line, and temperature. The MAX8664 operates with a constant switching frequency adjustable from 100kHz to 1MHz. Built-in boost diodes reduce external component count. Digital softstart eliminates input inrush current during startup. The second output has an optional REFIN2 input that takes an external reference voltage, facilitating tracking supply applications. Each output is capable of sourcing and sinking current. Internal 6.5V and 5V linear regulators provide power for gate drive and internal IC functions. The MAX8664 has built-in protection against output overvoltage, overcurrent, and thermal faults. The MAX8664B latches off both controllers during a fault condition, while the MAX8664A allows one controller to continue to function when there is a fault in the other controller. The MAX8664 employs Maxim's proprietary peak-voltage mode control architecture that provides superior transient response during either load or line transients. This architecture is easily stabilized using two resistors and one capacitor for any type of output capacitors. Fast transient response requires less output capacitance, consequently reducing total system cost.
Internal Linear Regulators
The internal VL low-dropout linear regulator of the MAX8664A and MAX8664B provides the 6.5V supply used for the gate drive. Connect a 4.7F ceramic capacitor from VL to PGND. When using a 4.5V to 5.5V input supply, connect VL directly to IN. The 5V supply used to power IC functions (VCC) is generated by an internal 1.5V shunt regulator from VL. Connect a 2.2F ceramic capacitor from VCC to GND. When using a 4.5V to 5.5V input supply, connect VCC to IN through a 10 resistor.
MAX8664
High-Side Gate-Drive Supply (BST_)
The gate-drive voltage for the high-side MOSFETs is generated using a flying capacitor boost circuit. The capacitor between BST_ and LX_ is charged to the VL voltage through the integrated BST_ diode during the low-side MOSFET on-time. When the low-side MOSFET is switched off, the BST_ voltage is shifted above the LX_ voltage to provide the necessary turn-on voltage (VGS) for the high-side MOSFET. The controller closes a switch between BST_ and DH_ to turn the high-side MOSFET on.
Voltage Reference
An internal 0.6V reference sets the feedback regulation voltage. Controller 1 always uses the internal reference. An external reference input is provided for controller 2. To use the external reference, connect a 0 to 1.3V supply to REFIN2. This facilitates tracking applications. To use the internal 0.6V reference for controller 2, connect REFIN2 to VCC.
Undervoltage Lockout (UVLO)
When the VCC supply voltage drops below the UVLO threshold (3.15V falling typ), the undervoltage lockout (UVLO) circuitry inhibits the switching of both controllers, and forces the DL and DH gate drivers low. When VCC rises above the UVLO threshold (3.5V rising typ), the controllers begin the startup sequence and resume normal operation.
DC-DC Controller Architecture
The peak-voltage mode PWM control scheme ensures stable operation, simple compensation for any output capacitor, and fast transient response. An on-chip integrator removes any DC error due to the ripple voltage. This control scheme is simple: when the output voltage falls below the regulation threshold, the error comparator begins a switching cycle by turning on the high-side switch at the rising edge of the following clock cycle. This switch remains on until the minimum on-time expires and the output voltage is in regulation or the current-limit threshold is exceeded. At this point, the low-side synchronous rectifier turns on and remains on until the rising edge of the first clock cycle after the output voltage falls below the regulation threshold.
Output Overcurrent Protection
When the MAX8664 detects an overcurrent condition, DH is immediately pulled low. If the overcurrent condition persists for four consecutive cycles, the controller latches off and both DH_ and DL_ are pulled low. During softstart, when FB_ is less than 300mV, the controller latches off on the first overcurrent condition. The protection circuit detects an overcurrent condition by sensing the drain-source voltage across the high-side MOSFET(s).
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
The threshold that trips overcurrent protection is set by a resistor connected from ILIM_ to the drain of the highside MOSFET(s). ILIM_ sinks 50A (typ) through this resistor. When the drain-source voltage exceeds the voltage drop across this resistor during the high-side MOSFET(s) on-time, an overcurrent fault is triggered. To prevent glitches from falsely tripping the overcurrent protection, connect a filter capacitor (0.01F typically) in parallel with the overcurrent-setting resistor.
Power-Good Output (PWRGD)
PWRGD is an open-drain output that is pulled low when the output voltage rises above the PWRGD upper threshold or falls below the PWRGD falling threshold. PWRGD is held low in shutdown, when VCC is below the UVLO threshold, during soft-start, and during fault conditions. PWRGD does not reflect the status of controller 2 in the MAX8664A, or when REFIN2 is connected to an external reference with either version. See Table 1 for PWRGD operation of the circuits of Figures 2-5 during fault conditions. For logic-level output voltages, connect an external pullup resistor between PWRGD and the logic power supply. A 100k resistor works well in most applications.
Output Overvoltage Protection (OVP)
During an overvoltage event on one or both of its outputs, the MAX8664 latches off the controller. This occurs when the feedback voltage exceeds its normal regulation voltage by 150mV for 10s. In this state, the low-side MOSFET(s) are on and the high-side MOSFET(s) are off to discharge the output. To clear the latch, cycle EN or the input power.
Fault-Shutdown Modes
When an overvoltage or overcurrent fault occurs on one controller of the MAX8664A, the second controller continues to operate. With the MAX8664B, a fault in one controller latches off both controllers automatically, and PWRGD is pulled low. See Table 1 for the fault-shutdown modes of the circuits shown in Figures 2-5.
Thermal-Overload Protection
Thermal-overload protection limits total power dissipation in the MAX8664. When the junction temperature exceeds +160C, an internal thermal sensor shuts down the device, pulling DH_ and DL_ low for both controllers. To restart the controller, cycle EN or input power.
Table 1. Fault Shutdown Modes for Circuits of Figures 2-5
CIRCUIT Figure 2, Figure 5 (Independent) Figure 3 (Tracking) Figure 4 (Sequenced) MAX8664A (INDEPENDENT) CONTROLLER 1 FAULT Controller 2 remains on. PWRGD is pulled low. Controller 2 shuts down. PWRGD is pulled low. Controller 2 shuts down. PWRGD is pulled low. CONTROLLER 2 FAULT Controller 1 remains on. PWRGD remains high. Controller 1 remains on. PWRGD remains high. Controller 1 remains on. PWRGD remains high. MAX8664B (JOINT) CONTROLLER 1 FAULT Controller 2 is shut down. PWRGD is pulled low. Controller 2 is shut down. PWRGD is pulled low. Controller 2 is shut down. PWRGD is pulled low. CONTROLLER 2 FAULT Controller 1 is shut down. PWRGD is pulled low. Controller 1 is shut down. PWRGD is pulled low. Controller 1 is shut down. PWRGD is pulled low.
10
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
C19 0.01F C20 10F N1 R3 51.1k C5 1500pF C1 10F C4 1000F
INPUT 10.8V TO 13.2V C17 1F C18 1F C14 4.7F
R1 2.7k FB1 IN DH1 VCC REFIN2 BST1 VL DL1 LX1 C13 0.22F N2 ILIM1
R4 3.92k
R5 1.15k
OUT1 2.5V/8A C23 0.1F
R37 3 C25 680pF
L1 1H
C6 47F
C7 47F
C8 47F
GND
MAX8664
PGND C16 0.01F
VCC R9 10k PWRGD R10 39.2k OSC/EN12 ENABLE ON OFF N9 2N7002 FB2 DH2 LX2 C15 0.22F BST2 DL2 N4 R38 3 C26 680pF R7 3.92k L2 1H R6 51.1k C12 1500F C9 47F C10 47F C11 47F OUT2 1.8V/8A C22 0.1F ILIM2 R2 3.01k N3 C21 10F C3 10F
POWER-GOOD TO SYSTEM
C27 0.47F
R8 1.82k
Figure 2. Low-Cost, 600kHz Typical Application Circuit
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
Table 2. Component List for Figure 2
DESIGNATION C1, C3, C20, C21 C4 QTY 4 DESCRIPTION 10F 20%, 16V X5R ceramic capacitors (1206) 1000F 20%, 16V electrolytic capacitor (8mm diameter, 20mm height) 1500pF, 50V C0G ceramic capacitors (0603) 47F 20%, 6.3V X5R ceramic capacitors (1206) 0.22F 10%, 25V X7R ceramic capacitors (0603) 4.7F 10%, 6.3V X5R ceramic capacitor (0805) 0.01F 10%, 50V X7R ceramic capacitors (0603) 1F 20%, 16V X5R ceramic capacitor (0603) 1F 20%, 6.3V X5R ceramic capacitor (0603) 0.1F 20%, 16V X7R ceramic capacitors (0603) DESIGNATION C25, C26 C27 L1, L2 N1-N4 N9 R1 R2 R3, R6 R4, R7 R5 R8 R9 R10 R37, R38 U1 QTY 2 1 2 4 1 1 1 2 2 1 1 1 1 2 1 DESCRIPTION 680pF, 50V C0G ceramic capacitors (0603) 0.47F 10%, 16V ceramic capacitor (0603) 1H inductors TOKO FDV0630-1R0M n-channel MOSFETs (8-pin SO) International Rectifier IRF7821 n-channel MOSFET (SOT23) Central 2N7002 2.74k 1% resistor (0603) 301k 1% resistor (0603) 51.1k 1% resistors (0603) 3.92k 1% resistors (0603) 1.15k 1% resistor (0603) 1.82k 1% resistor (0603) 10k 5% resistor (0603) 39.2k 1% resistor (0603) 3 5% resistors (0805) MAX8664 (20-pin QSOP)
1
C5, C12 C6-C11 C13, C15 C14 C16, C19 C17 C18 C22, C23
2 6 2 1 2 1 1 2
12
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
C1 0.01F C2 10F C3 10F C4 1000F
R1 3.16k FB1 INPUT 10V TO 14V C5 1F C6 1F REFIN2 R7 1k C12 1000pF VL C13 4.7F GND BST1 IN VCC LX1 C7 0.22F ILIM1 DH1 N1 N2
R2 24.3k
C8 0.015F
R3 10k
R4 3.57k
OUT1 R6 1k
L1 0.56H C9 470F
OUT1 1.8V/20A C10 470F C11 10F
R5 3 N3 N4 C14 2200pF
MAX8664
DL1
VCC
PGND C15 0.01F
R9 10k POWER-GOOD TO SYSTEM R10 44.2k OSC/EN12 ENABLE ON OFF FB2 LX2 N7 2N7002 BST2 DL2 N6 C18 0.22F PWRGD R8 2.74k ILIM2 DH2 N5
C16 10F
C17 10F
L2 0.47H R12 2 C23 2200pF R11 14.7k C19 4700pF C20 680F
OUT2 0.9V/6A C21 680F C22 10F
R13 10k
Figure 3. 500kHz Tracking Circuit for DDR2 Applications
______________________________________________________________________________________
13
Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
Table 3. Component List for Figure 3
DESIGNATION C1, C15 C2, C3, C16, C17 C4 C5 C6 C7, C18 C8 C9, C10 C11, C22 C12 C13 C14, C23 C19 C20, C21 L1 QTY 2 4 1 1 1 2 1 2 2 1 1 2 1 2 1 DESCRIPTION 0.01F, 10V X7R ceramic capacitors 10F, 16V X5R ceramic capacitors 1000F/16V aluminum electrolytic capacitor Rubycon 16MBZ1000M 1F, 16V X5R ceramic capacitor 1F, 10V X5R ceramic capacitor 0.22F, 10V X7R ceramic capacitors 0.015F, 10V X7R ceramic capacitor 470F, 2.5V POS capacitors Sanyo 2R5TPD470M6 10F, 6.3V X5R ceramic capacitors 1000pF, 10V X7R ceramic capacitor 4.7F, 10V X5R ceramic capacitor 2200pF, 25V X7R capacitors 4700pF, 10V X7R capacitor 680F, 2.5V POS capacitors Sanyo 2R5TPD680M6 0.56H, 4.6m inductor Panasonic ETQP4LR56WFL N6 N7 R1 R2 R3, R13 R4 R5 R6, R7 R8 R9 R10 R11 R12 1 1 1 1 2 1 1 2 1 1 1 1 1 DESIGNATION L2 N1, N2 N3, N4 N5 QTY 1 2 2 1 DESCRIPTION 0.47H, 1.2m inductor TOKO FDV0603-R47M n-channel MOSFETs IRLR7821 (D-Pak) n-channel MOSFETs IRLR3907Z (D-Pak) n-channel MOSFET IRF7807Z (8-pin SO) n-channel MOSFET IRF7821 (8-pin SO) n-channel MOSFET 2N7002 (SOT23) 3.16k 1% resistor (0402 or 0603) 24.3k 1% resistor (0402 or 0603) 10k 1% resistors (0402 or 0603) 3.57k 5% resistor (0402 or 0603) 3.0 5% resistor (0603) 1k 1% resistors (0402 or 0603) 2.74k 1% resistor (0402 or 0603) 10k 5% resistor (0402 or 0603) 44.2k 1% resistor (0402 or 0603) 14.7k 1% resistor (0402 or 0603) 2.0 5% resistor (0402 or 0603)
14
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
C1 0.01F C2 1F N1 IRF7821 R2 17.4k L1 0.2H R6 2 C13 2200pF C15 0.01F GND ILIM2 R11 3.32k R10 10k PWRGD R12 22.6k OSC/EN12 DH2 N3 IRF7821 LX2 L2 0.2H BST2 C19 0.22F R13 2 N4 IRF7821 C20 2200pF R14 17.4k C21 820pF C22 47F C23 47F OUT2 1.2V/10A C24 0.1F C16 1F C17 10F C18 10F INPUT 2.97V TO 3.63V C8 820pF R3 10k R4 3.16k OUT1 1.8V/10A C11 0.1F C3 10F C4 10F
R1 3.32k FB1 5V IN C5 1F 0.6V EXT REF VCC R7 10k C6 4.7F VL R5 10 VCC LX1 C7 0.22F BST1 N2 IRF7821 DH1 ILIM1
C12 1F
MAX8664
C9 47F
C10 47F
REFIN2 R8 10k C14 0.01F N5 2N7002 R9 47k POWER-GOOD TO SYSTEM Q1 CMST3904 VCC
DL1 PGND
FB2
DL2
R15 10k
R16 6.34k
Figure 4. 1MHz Application Circuit with All Ceramic Capacitors and Sequenced Outputs
______________________________________________________________________________________
15
Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
Table 4. Component List for Figure 4
DESIGNATION C1, C14, C15 C2, C16 C3, C4, C17, C18 C5, C12 C6 C7, C19 C8, C21 C9, C10, C22, C23 C11, C24 C13, C20 QTY 2 2 4 2 1 2 2 4 2 2 DESCRIPTION 0.01F, 10V X7R ceramic capacitors 1F, 6.3V X5R ceramic capacitors 10F, 6.3V X5R ceramic capacitors 1F, 10V X5R ceramic capacitors 4.7F, 10V X5R ceramic capacitor 0.22F, 10V X7R ceramic capacitors 820pF,10V X7R ceramic capacitors 47F, 6.3V X5R ceramic capacitors 0.1F, 10V X7R ceramic capacitors 2200pF, 25V X7R ceramic capacitors DESIGNATION L1, L2 N1-N4 N5 Q1 R1, R11 R2, R14 R3, R15 R4 R5 R6, R13 R7, R8, R10 R9 R12 R16 QTY 2 4 1 1 2 2 2 1 1 2 3 1 1 1 DESCRIPTION 0.2H, 2.4m inductors TOKO FDV0603-R20M n-channel MOSFETs IRF7821 (8-pin SO) n-channel MOSFET 2N7002 (SOT23) Transistor, bipolar, npn Central CMST3904 3.32k 1% resistors (0402 or 0603) 17.4k 1% resistors (0402 or 0603) 10k 1% resistors (0402 or 0603) 3.16k 1% resistor (0402 or 0603) 10.0 5% resistor (0402 or 0603) 2.0 5% resistors (0603) 10k 5% resistors (0402 or 0603) 47k 5% resistor (0402 or 0603) 22.6k 1% resistor (0402 or 0603) 6.34k 1% resistor (0402 or 0603)
16
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
C1 0.01F C2 10F N1 INPUT 7.2V TO 20V C5 1F IN DH1 LX1 VCC BST1 C7 0.22F N2 R2 40.2k L1 1.43H C9 470F C10 10F C8 4700pF R3 10k R4 5.36k OUT1 1.5V/10A C3 10F C4 OPEN
R1 2.87k FB1 ILIM1
C6 1F
MAX8664
REFIN2 C12 4.7F VL
DL1
R5 2 C11 1000pF
PGND C13 0.01F
GND ILIM2 VCC DH2 POWER-GOOD TO SYSTEM R7 10k PWRGD R8 75k OSC/EN12 ENABLE N5 2N7002 DL2 FB2 BST2 N4 LX2 C16 0.22F R6 2.26k N3
C14 10F
C15 10F
L2 1.43H C17 R9 25.5k 4700pF C18 470F C19 10F
OUT2 1.05V/8A
R10 2 C20 1000pF
R11 10k
R12 9.53k
Figure 5. 300kHz Circuit with 7.2V to 20V Input
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17
Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
Table 5. Component List for Figure 5
DESIGNATION C1, C13 C2, C3, C14, C15 C5 C6 C7, C16 C8, C17 C9, C18 C10, C19 C11, C20 C12 QTY 2 4 1 1 2 2 2 2 2 1 DESCRIPTION 0.01F, 10V X7R ceramic capacitors 10F, 25V X5R ceramic capacitors 1F, 25V X5R ceramic capacitor 1F, 10V X5R ceramic capacitor 0.22F, 10V X7R ceramic capacitors 4700pF, 10V X7R ceramic capacitors 470F/2.5V POSCAP capacitors Sanyo 2R5TPD470M6 10F, 6.3V X5R ceramic capacitors 1000pF, 25V X7R ceramic capacitors 4.7F, 10V X5R ceramic capacitor N5 R1 R2 R3, R11 R4 R5, R10 R6 R7 R8 R9 R12 1 1 1 2 1 2 1 1 1 1 1 DESIGNATION L1, L2 N1-N4 QTY 2 4 DESCRIPTION 1.43H, 4.52m inductors Panasonic ETQP3H1E4BFA n-channel MOSFETs IRF7821 (8-pin SOs) n-channel MOSFET 2N7002 (SOT23) 2.87k 1% resistor (0402 or 0603) 40.2k 1% resistor (0402 or 0603) 10k 1% resistors (0402 or 0603) 5.36k 1% resistor (0402 or 0603) 2.0 5% resistors (1206) 2.26k 1% resistor (0402 or 0603) 10k 5% resistor (0402 or 0603) 75k 1% resistor (0402 or 0603) 25.5k 1% resistor (0402 or 0603) 9.53k 1% resistor (0402 or 0603)
Power-Up and Sequencing
The MAX8664 features an OSC/EN12 input that is used both for setting the switching frequency and as an enable input for both controllers. A resistor from OSC/EN12 to GND sets the switching frequency, and when OSC/EN12 is high impedance, both controllers enter low-power shutdown mode. This is easily achieved with a transistor between the resistor and GND. Figure 6a shows the startup configuration with independent outputs. With REFIN2 connected to VCC, both controllers use the internal reference.
For tracking applications, connect REFIN2 to the center of a resistive voltage-divider between the output of controller 1 and GND. See Figure 6b. In this application, the output of regulator 2 tracks the output voltage of controller 1. The voltage-divider resistors set the VOUT2/VOUT1 ratio. A typical tracking application is for the VTT supply of DDR memory. Figure 6c shows one method of sequencing the outputs. Output 1 rises first. When PWRGD goes high, the transistors allow the external reference to drive REFIN2 and output 2 rises. The circuit in Figure 6d functions similarly, except the enable signal is supplied externally instead of being driven by the PWRGD signal.
CHIP ENABLE VCC VOUT1 REFIN2
VOUT2 ON PWRGD OFF CHIP ENABLE
MAX8664
OSC/EN12
Figure 6a. Two Independent Output Startup and Shutdown Waveforms
18 ______________________________________________________________________________________
Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
CHIP ENABLE VOUT1
VOUT1
REFIN2
VOUT2 ON PWRGD OFF CHIP ENABLE
MAX8664
OSC/EN12
Figure 6b. Ratiometric Tracking Startup and Shutdown Waveforms
VCC
EXTERNAL REF
CHIP ENABLE
PWRGD REFIN2
VOUT1
MAX8664
VOUT2 ON PWRGD OFF CHIP ENABLE OSC/EN12
Figure 6c. Sequencing Startup and Shutdown Waveforms
______________________________________________________________________________________
19
Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
VCC
CHIP ENABLE OUT2 ENABLE
EXTERNAL REF
REFIN2 ON OFF
VOUT1 OUT2 ENABLE VOUT2
MAX8664
ON OSC/EN12
PWRGD
OFF CHIP ENABLE
Figure 6d. Sequencing Startup and Shutdown Waveforms with System Enable 2 Signal
Design Procedure
Setting the Switching Frequency
Connect a resistor from OSC/EN12 to GND to set the switching frequency between 100kHz and 1000kHz. Calculate the resistor value (R10 in Figures 2-5) as follows: R10 = 2.24 x 1010 (Hz) () fS
Inductor Selection
There are several parameters that must be examined when determining which inductor is to be used. Input voltage, output voltage, load current, switching frequency, and LIR. LIR is the ratio of inductor-current ripple to maximum DC load current (ILOAD(MAX)). A higher LIR value allows for a smaller inductor, but results in higher losses and higher output ripple. A good compromise between size and efficiency is an LIR of 0.3. Once all the parameters are chosen, the inductor value is determined as follows: L= VOUT x (VIN - VOUT ) VIN x fS x ILOAD(MAX) x LIR
inductor value is not critical and can be adjusted to make trade-offs among size, cost, and efficiency. Lower inductor values minimize size and cost, but they also increase the output ripple and reduce the efficiency due to higher peak currents. On the other hand, higher inductor values increase efficiency, but eventually resistive losses due to extra turns of wire exceed the benefit gained from lower AC current levels. This is especially true if the inductance is increased without also increasing the physical size of the inductor. Find a low-loss inductor having the lowest possible DC resistance that fits the allotted dimensions. The chosen inductor's saturation current rating must exceed the peak inductor current determined as: IPEAK = ILOAD(MAX) + LIR x ILOAD(MAX) 2
Output Capacitor
The key selection parameters for the output capacitor are the actual capacitance value, the equivalent series resistance (ESR), the equivalent series inductance (ESL), and the voltage-rating requirements. These parameters affect the overall stability, output voltage ripple, and transient response. The output ripple has three components: variations in the charge stored in the output capacitor, the voltage drop across the capacitor's ESR, and ESL caused by the current into and out of the capacitor. The maximum output voltage ripple is estimated as follows:
where fS is the switching frequency. Choose a standard value inductor close to the calculated value. The exact
20
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response
VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) + VRIPPLE(ESL) The output voltage ripple as a consequence of the ESR, ESL, and output capacitance is: VRIPPLE(ESR) = IP-P x ESR VRIPPLE(ESL) = VRIPPLE(C) = VIN x ESL L + ESL
MAX8664
IP-P 8 x COUT x fS
output voltage instantly changes by ESR x ILOAD. Before the controller can respond, the output voltage deviates further depending on the inductor and output capacitor values. After a short period of time (see the Typical Operating Characteristics), the controller responds by regulating the output voltage back to its nominal state. The controller response time depends on its closed-loop bandwidth. With a higher bandwidth, the response time is faster, thus preventing the output voltage from further deviation from its regulating value.
Setting the Output Voltages and Voltage Positioning
Figure 7 shows the feedback network used on the MAX8664. With this configuration, a portion of the feedback signal is sensed on the switched side of the inductor (LX), and the output voltage droops slightly as the load current is increased due to the DC resistance of the inductor (DCR). This allows the load regulation to be set to match the voltage droop during a load transient (voltage positioning), reducing the peak-to-peak output voltage deviation during a load transient, and reducing the output capacitance requirements. To set the magnitude of the voltage positioning, select a value for R2 in the 8k to 24k range, then calculate the value of R1 as follows: IOUT(MAX) x DCR R1 = R2 x - 1 VOUT(MAX) where IOUT(MAX) is the maximum output current and VOUT(MAX) is the maximum allowable droop in the output voltage at full load.
where IP-P is the peak-to-peak inductor current: V -V V IP-P = IN OUT x OUT fS x L VIN These equations are suitable for initial capacitor selection, but final values should be chosen based on a prototype or evaluation circuit. As a general rule, a smaller ripple current results in less output-voltage ripple. Since the inductor ripple current is a factor of the inductor value and input voltage, the output-voltage ripple decreases with larger inductance, and increases with higher input voltages. Ceramic, tantalum, or aluminum polymer electrolytic capacitors are recommended. The aluminum electrolytic capacitor is the least expensive; however, it has higher ESR and ESL. To compensate for this, use a ceramic capacitor in parallel to reduce the switching ripple and noise. For reliable and safe operation, ensure that the capacitor's voltage and ripple-current ratings exceed the calculated values. The response to a load transient depends on the selected output capacitors. After a load transient, the
L LX_
DCR OUT
ESR R1 Cr R2 COUT
RLOAD
FB_ R3
Figure 7. Feedback Network
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
To set the no-load output voltage (VOUT), calculate the value of R3 as follows: R1 x R2 VFB R3 = VOUT - VFB R1 + R2 where VFB is the feedback regulation voltage (0.6V when using the internal reference or VREFIN2 for external reference). If the desired output voltage is equal to the reference voltage (typical for tracking applications), R3 is not installed. To achieve the lowest possible load regulation in applications where voltage positioning is not desired, R1 is not installed and R3 is calculated as follows: VFB R3 = x R2 VOUT - VFB Finally, calculate the value of Cr as follows: VOUT (VIN - VOUT ) VIN Cr = R1x fS x | (VFB _ RIPPLE - VOUT _ RIPPLE ) |
MOSFET Selection
Each output of the MAX8664 is capable of driving two to four external, logic-level, n-channel MOSFETs as the circuit switch elements. The key selection parameters are: * On-resistance (RDS(ON))--the lower, the better. * Maximum Drain-to-Source Voltage (VDSS)--should be at least 20% higher than the input supply rail at the high-side MOSFET's drain. Gate charges (Qg, Qgd, Qgs)-- the lower, the better.
*
Compensation
To ensure stable operation, connect a compensation capacitor (Cr) across the upper feedback resistor as shown in Figure 7. To find the value of this capacitor, follow the compensation design procedure below. Choose a closed-loop bandwidth (fC) that is less than 1/3 the switching frequency (fS). Calculate the output double pole (fO) as follows: fO = 1 R + ESR 2 L x COUT x LOAD RLOAD + DCR
For a 5V input application, choose MOSFETs with rated RDS(ON) at VGS 4.5V. With higher input voltages, the internal VL regulator provides 6.5V for gate drive in order to minimize the on-resistance for a wide range of MOSFETs. For a good compromise between efficiency and cost, choose the high-side MOSFETs that have conduction losses equal to switching losses at nominal input voltage and output current. Low RDS(ON) is preferred for lowside MOSFETs. Make sure that the low-side MOSFET(s) does not spuriously turn on due to dV/dt caused by the high-side MOSFET(s) turning on, as this would result in shoot-through current and degrade the efficiency. MOSFETs with a lower Q gd / Q gs ratio have higher immunity to dV/dt. For high-current applications, it is often preferable to parallel two MOSFETs rather than to use a single large MOSFET. For proper thermal management, the power dissipation must be calculated at the desired maximum operating junction temperature, maximum output current, and worst-case input voltage. For the-low side MOSFET(s), the worst-case power dissipation occurs at the highest duty cycle (VIN(MAX)). The low-side MOSFET(s) operate as zero voltage switches; therefore, major losses are the channel conduction loss (P LSCC) and the body diode conduction loss (PLSDC): VOUT 2 PLSCC(MAX) = 1 - xI x RDS(ON) VIN(MAX) LOAD(MAX) Use RDS(ON) at TJ(MAX): PLSDC(MAX) = 2 x ILOAD(MAX) VF x tDT x fS where VF is the body diode forward-voltage drop, tDT is the dead time between high-side and low-side switching transitions (25ns typical), and fS is the switching frequency.
The FB peak-to-peak voltage ripple is: R2 1+ VOUT R1 x VFB _ RIPPLE = R2 R2 DCR fC + 1+ R3 R1 1+ RLOAD x fO
The output ripple voltage due to the ESR of the output capacitor, COUT, is: VOUT (VIN - VOUT ) V VOUT _ RIPPLE = IN x L x fS 1 ESR + 8 x CO x fS Target the feedback ripple in the 25mV to 60mV range. For high duty-cycle applications (> 70%), a feedback ripple of 25mV is recommended.
22
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response
The high-side MOSFET(s) operate as duty-cycle control switches and have the following major losses: the channel conduction loss (PHSCC), the overlapping switching loss (PHSSW), and the drive loss (PHSDR). The maximum power dissipation could occur either at VIN(MAX) or VIN(MIN): PHSCC(MAX) = VOUT x I2 x RDS(ON) LOAD(MAX) VIN(MIN) interfere with circuit performance and generate EMI. To dampen this ringing, a series RC snubber circuit is added across each low-side switch. Below is the procedure for selecting the value of the series RC circuit. Connect a scope probe to measure VLX_ to GND and observe the ringing frequency, fR. Find the capacitor value (connected from LX_ to GND) that reduces the ringing frequency by half. The circuit parasitic capacitance (CPAR) at LX_ is then equal to 1/3 the value of the added capacitance above. The circuit parasitic inductance (LPAR) is calculated by: LPAR = 1
MAX8664
Use RDS(ON) at TJ(MAX): PHSSW(MAX) = VIN(MAX) x ILOAD(MAX) x QGD x fS IGATE
(2fR )
2
x CPAR
where IGATE is the average DH driver output-current capability determined by: IGATE 0.5 x VVL RDS(ON)(DR) + RGATE
where RDS(ON)(DR) is the DH_ driver's on-resistance (see the Electrical Characteristics) and RGATE is the internal gate resistance of the MOSFET (~ 2): PHSDR = QG x VGS x fS x RGATE RGATE + RDS(ON)(DR)
The resistor for critical dampening (RSNUB) is equal to 2 x fR x LPAR. Adjust the resistor value up or down to tailor the desired damping and the peak-voltage excursion. The capacitor (CSNUB) should be at least 2 to 4 times the value of the CPAR to be effective. The power loss of the snubber circuit is dissipated in the resistor (PRSNUB) and can be calculated as: PRSNUB = CSNUB x (VIN ) x fSW
2
where VGS VVL. The high-side MOSFET(s) do not have body diode conduction loss, unless the converter is sinking current. When sinking current, calculate this loss as PHSDC(MAX) = ILOAD(MAX) x VF x (2 x tDT + tWD) x fS, where tWD is about 130ns. Allow an additional 20% for losses due to MOSFET output capacitances and low-side MOSFET body diode reverse-recovery charge dissipated in the high-side MOSFET(s). Refer to the MOSFET data sheet for thermal resistance specifications to calculate the PCB area needed to maintain the desired maximum operating junction temperature with the above calculated power dissipations.
where VIN is the input voltage and fSW is the switching frequency. Choose an RSNUB power rating that meets the specific application's derating rule for the power dissipation calculated.
Setting the Overcurrent Threshold
Connect a resistor from ILIM_ to the drain of the highside MOSFET(s) to set the overcurrent protection threshold. ILIM_ sinks 50A (typ) through this resistor. When the drain-source voltage exceeds the voltage drop across this resistor during the high-side MOSFET(s) on-time, overcurrent protection is triggered. To set the output current level where overcurrent protection is triggered (ILIMIT), calculate the value of the ILIM_ resistor as follows: R ILIM _ = RDS(ON)HS x ILIMIT 50A
MOSFET Snubber Circuit
Fast switching transitions cause ringing because of resonating circuit parasitic inductance and capacitance at the switching nodes. This high-frequency ringing occurs at LX's rising and falling transitions and can
where RDS(ON)HS is the maximum on-resistance of the high-side MOSFET(s) at +25C. At higher temperatures, the ILIM current increases to compensate for the temperature coefficient of the high-side MOSFET(s).
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23
Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
Input Capacitor
The input filter capacitors reduce peak currents drawn from the power source and reduce noise and voltage ripple on the input caused by the circuit's switching. The input capacitors must meet the ripple current requirement (IRMS) imposed by the switching currents. The ripple current requirement can be estimated by the following equation:
IRMS = 1 VIN
way that the high-side MOSFET's drain is close and near the low-side MOSFET's source. This allows the input ceramic decoupling capacitor to be placed directly across and as close as possible to the high-MOSFET's drain and the low-side MOSFET's source. This helps contain the high switching current within this small loop. 3) Pour an analog ground plane in the second layer underneath the IC to minimize noise coupling. 4) Connect input, output, and VL capacitors to the power ground plane; connect all other capacitors to the signal ground plane. 5) Place the MOSFETs as close as possible to the IC to minimize trace inductance of the gate drive loop. If parallel MOSFETs are used, keep the trace lengths to both gates equal and short. 6) Connect the drain leads of the power MOSFET to a large copper area to help cool the device. Refer to the power MOSFET data sheet for recommended copper area. 7) Place the feedback network components as close as possible to the IC pins. 8) The current-limit setting RC should be Kelvin connected to the high-side MOSFETs' drain. Refer to the MAX8664 evaluation kit for an example layout.
(IOUT1)2 x VOUT1 x (VIN - VOUT1) + (IOUT2 )2 x VOUT2 x (VIN - VOUT2 )
Choose a capacitor that exhibits less than 10C temperature rise at the maximum operating RMS current for optimum long-term reliability.
Applications Information
PCB Layout Guidelines
Careful PCB layout is an important factor in achieving low switching losses and clean, stable operation. The switching power stage requires particular attention. Follow these guidelines for good PCB layout: 1) A multilayer PCB is recommended. 2) Place IC decoupling capacitors as close as possible to the IC pins. Keep separate power ground and signal ground planes. Place the low-side MOSFETs near the PGND pin. Arrange the highside MOSFETs and low-side MOSFETs in such a
24
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Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response
Pin Configuration
PROCESS: BiCMOS
TOP VIEW
DH1 1 LX1 2 BST1 3 DL1 4 VL 5 PGND 6 DL2 7 BST2 8 LX2 9 DH2 10 20 ILIM1 19 FB1 18 PWRGD
Chip Information
MAX8664
MAX8664
17 VCC 16 GND 15 IN 14 OSC/EN12 13 REFIN2 12 FB2 11 ILIM2
QSOP
______________________________________________________________________________________
25
Low-Cost, Dual-Output, Step-Down Controller with Fast Transient Response MAX8664
Package Information (continued)
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information go to www.maxim-ic.com/packages.)
QSOP.EPS
PACKAGE OUTLINE, QSOP .150", .025" LEAD PITCH
21-0055
F
1 1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
26 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2007 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.

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